Integrated Broadband Coding Metasurface for Simultaneous Tunable Radiation and Low‑Scattering Performance

Background

Electromagnetic metasurfaces (EMMSs) are engineered two‑dimensional arrays of sub‑wavelength resonators that control phase, amplitude, and polarization of incident waves [1,2]. Their thin profile, conformability, and low‑cost fabrication have driven extensive research into wavefront shaping, polarization conversion, and absorption [3–9]. Anisotropic particles further expand the design space, enabling functions unattainable with isotropic elements, such as arbitrary linear‑to‑linear or linear‑to‑circular polarization conversion [10–18], perfect absorption through near‑field coupling [19–21], and high‑performance phase discontinuities for lenses, holograms, and cloaks [22–31].

Recently, coding EMMSs have emerged as a powerful paradigm for EM wave manipulation. Binary or multibit coding assigns discrete phase states (e.g., 0° and 180° for 1‑bit) to unit cells, allowing arbitrary spatial phase patterns [32–38]. Incorporating reconfigurable or programmable elements further extends these metasurfaces to tunable and adaptive applications [39–40]. Although most coding EMMS designs focus on scattering control, their intrinsic radiative capabilities have been largely overlooked.

Our work bridges this gap by designing an anisotropic coding metasurface that simultaneously delivers broadband low‑scattering performance and tunable radiation. By carefully selecting unit‑cell geometry, we achieve complementary phase responses for orthogonal polarizations, enabling the metasurface to serve as an efficient antenna when appropriately fed.

Methods

Figure 1 illustrates the unit‑cell geometry and the resulting 4 × 4 coding metasurface. Two FR‑4 dielectric layers (εr = 2.65, tan δ = 0.002) with thicknesses 3 mm and 0.5 mm are stacked without air gaps. Four 4 × 4 bowtie patches (36 mm × 36 mm) are etched on the top layer, and a narrow‑slot ground plane (15.5 mm × 0.2 mm) is etched on the bottom layer to enforce near‑perfect reflection. The “1” cell (Fig. 1b) and its 90° rotated counterpart “0” (Fig. 1c) exhibit opposite phase responses under orthogonal polarization. The optimal 0/1 layout is obtained via a simulated‑annealing algorithm (SAA) that minimizes the peak array factor, thereby maximizing diffusion scattering. Key SAA parameters are initial temperature = 100, cooling factor = 0.9, final temperature = 0, and 1000 iterations.

Coding EMMS and its constituent anisotropic element. a 4 × 4 array with equal numbers of “0” and “1”. b “1” element geometry; c “0” element geometry. d SAA flowchart.

For radiation simulations, each cell is excited via a lumped port and a 50 Ω SMA connects to a narrow rectangular patch (13 mm × 1.3 mm) through a hole in the bottom substrate to provide impedance matching. A thin slot in the ground plane couples energy into the top layer, enabling broadside radiation. Figure 2 shows a 1‑dB bandwidth from 5 GHz to 6 GHz (18.2% relative bandwidth) and a stable boresight gain of 7 dBi. Radiation patterns in the E‑ and H‑planes are symmetric and broadside‑directed.

Radiation properties of the anisotropic element. a S11 and boresight gain versus frequency. b–c 2‑D patterns in E‑ and H‑planes. d 3‑D patterns at 5.35, 5.5, and 5.75 GHz.

Modal analysis (FEKO 7.0) reveals that the first two orthogonal modes (mode 1 at 5.32 GHz, mode 2 at 5.72 GHz) dominate the broadband response, while higher‑order modes are suppressed. Figure 3 visualizes surface currents for modes 1–4, illustrating the fundamental orthogonal mode pair responsible for the desired radiation.

Surface currents of modes 1–4 at 5.35 GHz (a) and 5.75 GHz (b).

Modal significance with (a) and without (b) bowtie metasurface.

Scattering performance is evaluated using Floquet ports. Reflection phase plots (Fig. 5a) show a 180° phase difference between “0” and “1” cells over 2–18 GHz, while reflection magnitudes remain near unity (Fig. 5b). This phase contrast ensures broadband energy cancellation, yielding low radar cross section (RCS). Full‑wave simulations confirm a 6‑dB RCS reduction from 5 GHz to 18 GHz, with two dips at 5.9 GHz and 10.4 GHz achieving up to 31.8 dB RCS suppression.

Reflection characteristics of the anisotropic element. a Phases; b Magnitudes.

Results and Discussion

The behavior of the coding metasurface can be understood via conventional array theory:

\[\mathbf{E}_{\text{total}}=\sum_{m=0}^{M-1}\sum_{n=0}^{N-1}\mathbf{EP}_{m,n}\,e^{j(k m\Delta x\sin\theta\cos\varphi + k n\Delta y\sin\theta\sin\varphi + \phi_{m,n})}\]

For radiation, equal‑amplitude feeds produce orthogonal field vectors; the relative feed phase controls the output polarization: 0° (or 180°) yields LP, while a 90° offset generates RHCP. Two representative operating points are presented.

With matched amplitude and phase (ϕ₀ = ϕ₁ = 0°), the array exhibits LP radiation across 4.97–6.05 GHz (19.6% bandwidth) and boresight gain from 12.6 to 17.4 dBi. Patterns remain symmetric in both planes (Fig. 6).

Linear radiation performance. a S11 and boresight gain. b 3‑D LP patterns at 5.35, 5.5, and 5.75 GHz.

For RHCP, a 90° phase difference between feeds yields a 3‑dB axial‑ratio bandwidth of 5.22–6.00 GHz (13.9% bandwidth) and boresight gain of 13.2–15.8 dBi (Fig. 7).

RHCP performance. a S11 and axial ratio. b Gain. c 3‑D RHCP patterns.

Electric‑field snapshots at 5.35 GHz (Fig. 8) confirm that LP radiation arises from in‑phase excitation of both cell types, whereas CP radiation results from a 90° phase shift that preferentially excites one cell type at a time.

Field distributions. a LP case. b RHCP case.

Scattering results (Fig. 9) show a continuous 6‑dB RCS reduction across 5–18 GHz, with two pronounced dips reaching 31.8 dB. The pattern evolution (Fig. 9c–e) demonstrates eight weak lobes instead of the four dominant lobes of a conventional chessboard array, confirming effective diffusion scattering. Oblique‑incidence simulations (Fig. 10) and measured scattering (Fig. 11) further verify broadband low‑scattering performance across angles up to 60°.

Diffusion scattering. a RCS vs frequency. b–c Calculated patterns. d–e Full‑wave patterns. f Surface currents.

Oblique‑incidence scattering patterns at 6 GHz.

Normalized scattering under varying incident angles.

The fabricated 4 × 4 metasurface was tested in an anechoic chamber. Measured S11 and axial‑ratio bandwidths (Fig. 13a) align closely with simulations, achieving 5.22–6.02 GHz (14.2% bandwidth). Radiation patterns (Fig. 13b–c) confirm symmetric, broadside, RHCP radiation with sidelobes at least 10 dB below the main lobe. Scattering measurements (Fig. 13d) demonstrate a 6‑dB RCS reduction from 5 to 18 GHz, with peaks at 6.1 GHz and 10.2 GHz reaching 26 dB and 31 dB respectively.

Fabrication and measurement setup. a–b Top and side views. c Power divider. d Scattering measurement layout.

Measured performance. a S11 and AR. b–c Radiation patterns. d RCS reduction.

Conclusions

We have demonstrated a coding EMMS that simultaneously delivers broadband, tunable radiation and broadband low‑scattering performance. By exploiting anisotropic unit cells with orthogonal phase responses and optimizing their binary layout via simulated annealing, the metasurface operates as an efficient antenna (LP or CP) while achieving a 6‑dB RCS reduction from 5 to 18 GHz. This dual‑functionality offers a straightforward, scalable platform for polarization‑reconfigurable antennas, stealth applications, and beyond.

Abbreviations

ARBW:

Axial ratio bandwidth

EM:

Electromagnetic

EMMS:

Electromagnetic metasurfaces

L/RHCP:

Left‑ or right‑hand circular polarization

LP:

Linearly polarized

PCB:

Printed circuit board

RCSR:

Radar cross section reduction

SAA:

Simulated annealing algorithm